Electronically reconfigurable, or electro-modulated microwave filters are attracting increasing attention for research and development due to their increasing importance in improving the capacity of present and future microwave systems. For example, the emergence of ultra-wideband (UWB) technology requires the use of a very wide radio spectrum. However, spectrum as a resource is valuable and finite, so it is always used for multiple purposes, which means that when operations such as UWB wireless systems are of interest, the spectrum is filled with undesired signals. In this case, the existing time-varying undesired narrowband radio signals have the potential to interfere with the band of the UWB system. The solution to this problem is to introduce a narrow rejection band (trap) on the passband of the UWB bandpass filter that can be electrically switched or electrically tuned. Such electronically reconfigurable filters are also desirable for wideband radar or electronic military systems. We can be proactive in considering future cognitive radio and radar applications, and it is certain that electronically reconfigurable microwave filters will play a more important role in wireless systems.
In general, in order to develop electronically reconfigurable filters, active switching or tuning elements such as semiconductor p-i-n and varactor diodes, radio frequency (RF) microelectromechanical systems (MEMS) or other functional material-based components, including ferroelectric varactors, need to be integrated into the passive filter structure. There is a growing interest in developing tunable or reconfigurable filters based on microstrip lines [2]-[36], since microstrip line filters [1] can facilitate such integration with very small dimensions. These filters can be classified as tunable comb bandpass filters [2]-[9], tunable filters for RF MEMS [10]-[15], tunable filters for piezoelectric sensors (PET) [17]-[19], tunable high-temperature superconducting (HTS) filters [21]-[23], reconfigurable UWB filters [24]-25], tunable dual-band filters [ 26], tunable bandstop filters [27]-[31], reconfigurable/tunable dual-mode filters [32]-[36], and reconfigurable bandpass filters based on switchable delay lines. In the following, we will present several newly developed typical electronically reconfigurable microstrip line filters.
Tunable comb filters
Microstrip line comb filters are a popular structure for developing tunable or reconfigurable bandpass filters [2]-[9]. Figure 1 shows a schematic diagram of a 3-pole tunable comb filter, where each microstrip line resonator with a length less than a quarter wavelength of the operating frequency is short-circuited at one end and loaded with a varactor at the other end. In this example, the varactors are based on barium strontium titanate (BST) thin films of ferroelectrics. The bias network of each BST varactor contains a spaced capacitor connected in series with the varactor. The center frequency of the bandpass filter can be electronically tuned by varying the dc bias applied to the varactor.
Figure 1, a schematic diagram of a tunable comb bandpass filter [2].
Figure 2 gives an example of the fabrication of a tunable comb filter monolithically integrated with a BST film and its measurement performance. As reported in the literature [3], for dc bias, a resistive tantalum nitride (TaN) film is deposited and patterned on top of the BST film, which has a surface resistivity of approximately 1-10 KΩ per unit area (Square) and is used to direct the dc bias signal to the circuit while minimizing the impact on the circuit RF performance. The integration of BST thin film capacitors with bauxite substrates, copper electrodes, and vias advances tunable dielectric technology in several ways, and this tuning technology allows the development of more complex RF and microwave circuits.
Figure 2, (a) an X-band (8-12 GHz) comb tunable filter employing a barium strontium titanate film and (b) its tested performance [3].
The bandwidth fluctuations that occur when performing center frequency tuning are a well known problem. In general, in order to maintain an absolute passband bandwidth independent of the tuning frequency, the coupling coefficient must be inversely proportional to the tuning frequency. Several techniques exist to solve this problem, such as in the literature [4] and [8]. In the literature [8], one investigates variable vessel tuned comb bandpass filters using stepped impedance microstrip line resonators, which allows better control of the coupling between resonators and thus allows the constant bandwidth requirement to be met by using shorter electrical length line elements. Also, aggregate inductors have been used to act as input and output coupling networks, thus allowing the external quality factor (Q) to vary directly with the tuning frequency. A quadrupole filter of this type has been demonstrated with a 3-dB passband bandwidth variation of less than 3.2% over a tuning range of 250 MHz at 2 GHz.
In the literature [4], theoretical analysis shows that for a tunable N-pole filter, N is the number of resonators and its quality factor (figure of merit) is defined as the ratio of the shift of the passband center frequency to the average passband, which depends on the loss or the Q value of the tuned varactor and the order of the filter. Since Q is inversely proportional to the power loss, the higher the loss, the lower the Q value. In general, the quality factor (or tuning range) is reduced for small Q values and large N values.
One can design tunable planar comb filters with multiple transmission zeros to improve the upper stopband performance by implementing source/load multiple resonator coupling. As demonstrated in the literature [9], this can be achieved by adding two new coupling lines to the classical comb configuration, as shown in Figure 3(a), where we can see two thin lines reaching from the source/load side to the internal resonator. Figure 3(b) plots the measured performance of the filter with five transmission zeros distributed in the upper resistance band. The center frequency of the filter can be tuned from 750 MHz to 900 MHz. The filter uses a Philips BB149 varactor diode, and the capacitance of this diode varies from 2 to 20 pF at a bias voltage of 0 V to 20 V. The biasing elements of the varactor are 6.8 pF and 22 nH.
Figure 3, (a) the fabricated 3-pole electro-modulated comb filter using source/load multiresonator coupling and (b) its measured response [9].
RF MEMS reconfigurable filter
A second class of reconfigurable filters is formed by using RF MEMS reconfigurable filters instead of varactors to vary the length of resonators or their circuit parameters in resonator filters [10]-[15]. In this case, electronic tuning is usually achieved digitally, which allows to achieve a large tuning range with good performance, which incorporates low losses and high linearity. Figure 4 illustrates an example of this type of filter. In this case, as reported in the literature [11], this topology is based on a distributed half-wavelength microstrip line resonator which is fabricated on a highly resistive silicon substrate. This is a bandpass filter with two resonators, and the addition of a capacitive patch at the end of each resonator allows for a low-loss shift of the pseudo 2-bits center frequency. The measured filter response is shown in Figure 4(b), where the passband can be reconstructed at four different center frequencies.
Figure 4, (a) Photograph of a 2-bit RF MEMS reconfigurable filter and (b) its measured response [11].
The pioneering work carried out on the differential 4-bit RF MEMS reconfigurable filter is reported in the literature [13]. This filter exhibits extremely fine tuning accuracy and can be filtered over a wide tuning range from 6.5 to 10 GHz, with different filter responses adjacent to each other at 16 frequencies, similar to a continuously tunable filter. To learn more about reconfigurable filters for RF MEMS, please refer to the article in this journal that focuses on this topic [16].
Tunable filters for piezoelectric sensors
Piezoelectric sensors (PETs) have been similarly used to develop electro-tunable microstrip line filters [17]-[19]. The construction of a PET tunable microstrip line filter is illustrated in Figure 5. As reported in the literature [17], this tunable filter circuit consists of a filter using a cascaded microstrip line open-loop resonator [20], a PET and a dielectric perturber attached to the top of the filter. The PET shown in Figure 5 is composed of two piezoelectric layers and a bedding layer. The spacer layer sandwiched between the two equally polarized piezoelectric layers adds mechanical strength and rigidity. The spacer is connected to an electrode of DC voltage to deflect the PET and allow vertical movement up and down. As we can see in the structure of Figure 5, the effective dielectric constant of the filter decreases or increases as the perturbers move up and down, respectively, causing the filter passband to shift to higher or lower frequencies.
Figure 5, Construction of the tunable bandpass filter [17]. (a) Top view. (b) Three-dimensional view.
Reconfigurable UWB filter
UWB filter with switchable trap
Figure 6 is a picture of the developed UWB bandpass filter with switchable trap bands [24]. Essentially, the UWB filter without trapping is an optimal distributed high-pass filter that contains five short-circuit cutoffs and four connecting line segments on the microstrip line [1].
Figure 6, Microstrip line UWB filter with two switchable trap structures [24].
In order to implement a switchable trap on the passband of the UWB filter, as shown in Figure 7. Two identical switchable trap structures are integrated into the two connecting line segments. A broadband DC bias circuit is added to this structure. In principle, the switchable trap structure in Fig. 7 is a section of transmission line with a mosaic cutoff [39]. When the p-i-n diode is at zero bias, it exhibits a large impedance due to its very small junction capacitance, so the mosaic cutoff acts as an open circuit cutoff that can generate resonance. Thus, at its fundamental frequency resonance frequency, the inlaid open-circuit cutoff shows a short circuit on the main transmission line, resulting in a narrow trap band or trap in the frequency response. This situation corresponds to a trap-on state. To turn off the trap, a forward bias is applied to the p-i-n diode. With a forward bias, the p-i-n diode corresponds to a very small resistance. Therefore, the open end of the mosaic intercept is connected to the main transmission line, so that there is no resonance from this mosaic intercept. As a result, the trapped wave disappears.
Figure 7, a schematic of a switchable trap structure [24].
Figure 8 shows the simulated and measured response of the reconfigurable UWB filter, where we can observe the on/off of the trap at a center frequency of about 5.1 GHz, with a rejection ratio greater than 35 dB when it is on. p-i-n diode (M/A-COM MA 4AGSBP907) is at zero bias in order to turn on the trap. When the trap is turned off, the performance of the filter is almost the same as when the p-i-n diode is forward biased at 2.5-5mA. The measured minimum insertion loss is 0.5 dB, and the measured 3 dB bandwidth is 5.92 GHz. The small difference between the simulated and measured results can be explained by manufacturing tolerances, p-i-n diodes, chip inductance, or spurious effects of capacitance.
Figure 8, Comparison of simulation and measurement results [24]. (Electromagnetic (EM) simulations were obtained using a commercially available tool [40].)
UWB filter with tunable trap
In Figure 6, one can generate a tunable trap structure by replacing the p-i-n diode with a varactor diode. Thus, the reconfigurable UWB filter discussed earlier can be modified to have an electronically tunable trap band. A GaAs flip-chip varactor diode with constant gamma from M/A-COM is used [25]. Figure 9(a) shows a typical performance curve of the MA46H120 varactor diode. For the experimental demonstration, the UWB filter with tunable trap was implemented on a liquid crystal polymer (LCD) with a substrate having a relative permittivity of 3.0 and a thickness of 0.5 mm. figure 9(b) shows the fabricated filter. The layout design is similar to that of Fig. 6 except that a varactor diode is used to replace the p-i-n diode. The varactor diode is connected to the DC voltage through a 10 kΩ resistor.
Figure 9, (a) Typical performance of MA46H120 varactor diode. (b) Reconfigurable ultra-wideband filter with tunable traps fabricated using the MA46H120 varactor diode [25].
Figure 10 shows the measured response of the reconfigurable UWB filter. When there is no DC bias (0V), there is no trap on the passband, as shown in Figure 10(a). This is because the varactor diode capacitance is so large at 0-V bias that it shifts the trap band below the passband. With a DC bias of 4V to 14V, the trap frequency is tuned in the 4.5GHz to 6.5GHz range in the UWB passband, as shown in Figure 10(b).
Figure 10. Measured performance of a reconfigurable ultra-wideband filter with tunable trap bands. (a) 0-V bias without trapping. (b) Non-zero bias with tunable trap (from [25]).
BST varactor-tuned bandstop filter with slotted wire ground structure
Ferroelectric materials such as BSTs are newly becoming more attractive in the development of electrically tuned microwave circuits for frequency shortcut applications [41]-[42]. Next, we would like to introduce the newly developed BST varactor tunable bandstop filter [28].
BST varactor
The capacitance tunable capability of the BST varactor can be defined as
Capacitance tunable capability = 3f37b3d6-a826-11ed-bfe3-dac502259ad0.jpg (1)
where Cmax is the capacitance of the BST varactor at 0-V bias and Cmin is the capacitance obtained at a non-zero DC bias. Cmin decreases when the absolute value of the dc bias voltage is increased because the relative dielectric constant of the BST material decreases with the applied voltage [42].
In general, BST varactors can be designed in the form of metal-insulator-metal capacitors or a form of fork-finger capacitor (IDC).BST tunable IDCs have a small capacitive tunability due to their additional edge capacitance that is not very sensitive to the dc voltage. Figure 11 illustrates the typical characteristics of BST IDCs [2], which are a more attractive choice when lower capacitance values and a simpler manufacturing process are required.
Figure 11, nominal tuning curve of the barium strontium titanate fork-finger capacitor varactor at 1 MHz (with 20 fork fingers, each fork finger is 5 μm wide and 100 μm long, and the fork finger spacing is 5 μm) [2].
As published in the literature [28], the IDC BST varactor was fabricated on the basis of the structure shown in Figure 12(a). A 0.5-μm-thick layer of Ba0.5Sr0.5TiO3 (BST) film was deposited onto a (001) MgO substrate (0.5-mm thickness) by a pulsed laser deposition method, using a laser with 1.5 Jcm-2 at an injection pulse rate of 5 Hz in an aerobic environment (0.1 mbar) [42]. The relative permittivity of the fabricated BST material varies from 700 to 1,200 as the electric field strength varies from 3.5 to 0 V/μm, and the measured dielectric loss tanδ of the BST film varies from 0.1 to 0.2 at 10 MHz.
Figure 12(b) shows the IDC with six forked fingers, which is the basic unit of the BST varactor developed in the literature [28]. The forked fingers spaced 10 μm apart from each other are 220-μm long and 10-μm wide. This basic BST varactor cell has Cmax=0.56pF at 0-V bias and Cmin=0.4pF at 35-V bias, and its capacitance tunability in a given DC bias voltage range is 28.6% according to equation (1). For practical applications, a large BST varactor chip has been made, and this chip can be easily attached to a conventional microwave circuit board. Its size is 5 × 5 mm2, containing three parallel BST varactor cells.
Figure 12, (a) Profile layer of the strontium barium titanate fork-finger transformer chip fabricated on MgO substrate. (b) Plate view of the strontium titanate barium fork-finger varactor cell (dimensions in mm) [data from 28].
Tunable bandstop filter
Figure 13 illustrates a two-pole tunable bandstop filter with a slotted line grounding structure. In the ground plane, the tunable microstrip line bandstop filter contains two tunable BST slotline resonators and a bias circuit that supplies DC voltage to the BST varactor.
Figure 13, Fabricated tunable microstrip line bandstop filter using strontium titanate barium varactor [28]. (a) Bottom and (b) top.
Figure 14 gives the measured response of the tunable microstrip line bandstop filter; this bandstop filter operates at a center frequency of 1.2-1.4 GHz and has a bandwidth of 100 MHz. Therefore, the measured tuning range is 14%.
Figure 14, Measurement results of the tunable bandstop filter [28].
Reconfigurable dual-mode bandpass filters
One can design bandpass filters based on single-mode or dual-mode resonators and prefers to do so on microstrip lines because the DC bias circuit can be easily implemented on microstrip lines. Dual-mode microstrip line resonators are attractive because each dual-mode resonator can be used as a dual-tuned resonant circuit, so that the number of resonators required for a given order of filter can be halved, resulting in a compact filter architecture. For a conventional dual-mode filter, the two simple modes are coupled by controlling a suitable perturbation. This is the case of the circular ring filter [43], the square loop filter, and the meander loop filter [45]. On the other hand, a novel dual-mode resonator filter based on a triangular patch has been studied in the literature [46], where no coupling of the dual modes is performed. Recently, this unique and interesting property has been demonstrated in a small dual-mode microstrip line open-loop filter [47], which evolved from a conventional single-mode open-loop resonator [20].
When a conventional dual-mode resonator with two-dimensional symmetry is used in the design of a dual-mode bandpass filter, some perturbation is required to separate the two simplex modes [1]. The dual-mode resonators presented in [46] and [47] do not require this perturbation because their two resonant modes, called even-mode and odd-mode, are not coupled to each other. These two modes will operate at separate frequencies in the dual-mode filter, and their coupling structure is different from that of the transmitted dual-mode filter. Figure 15 illustrates the coupling structure of a two-pole dual-mode filter of this type, where S and L represent the input and output ports, respectively; node 1 represents the odd mode and node 2 the even mode. This type of filter with a fixed center frequency is demonstrated in the literature [46] and [47].
Figure 15, Coupling structure of a two-pole dual-mode filter in which the two modes are uncoupled from each other [35].
In the literature [34]-[36], one similarly investigates electronically reconfigurable dual-mode microstrip line open-loop resonator filters that uncover the unique property of no coupling between the two resonant modes in a single dual-mode resonator. This results in a simple tuning scheme, since the tuning of the passband frequency can be accomplished simply by varying the mode frequency proportionally. In addition, for this type of filter, the selectivity can be electronically reset so that either side of the passband exhibits a higher selectivity with a finite frequency transmission zero point.
Reconfigurable dual-mode filter with two dc biases
As discussed in the literature [34], Figure 16 shows the fabricated two-pole reconfigurable dual-mode microstrip line open-loop resonator bandpass filter.
Figure 16, fabricated reconfigurable dual-mode microstrip line open-loop resonator bandpass filter with two DC biases [34].
This bandpass filter has a coupling scheme as shown in Figure 15. Since there is no coupling between the two operating resonant modes, it is possible to tune the center frequency of the passband if the resonant frequencies of the odd and even modes are shifted proportionally. The Infineon BB857 varactor with a typical variable capacitance varying between 0.5pF and 6.6pF is used to implement this electronic tuning. To be able to reconfigure the filter characteristics, two DC biases are used. The first DC bias voltage V1 was used to vary the odd-mode frequency and the second DC bias voltage V2 was used to vary the even-mode frequency.
The measured frequency response is plotted in Figure 17, which shows that, depending on the combination of the two dc biases, one can not only tune the passband frequency, but also reconfigure the filter characteristics, changing from a high selectivity on the high frequency side of the passband in the first case to a high selectivity on the low frequency side of the passband in the second case.
Figure 17, Measured performance of a reconfigurable dual-mode filter with two DC biases. (a) The first case. (b) The second case [34].
Reconfigurable dual-mode filter with a single dc bias
By varying the dual-mode microstrip line open-loop resonator, the center frequency of the dual-mode filter can be electronically tuned by employing a single dc bias. In other words, the change of resonant frequencies for even and odd modes is made easier by using the same bias voltage. Figure 18(a) shows an example of such a tunable filter [35]. This filter was fabricated on a base with a relative permittivity of 10.8 and a thickness of 1.27 mm. Three Infineon BB857 varactors are connected to this filter, which is similar to the previous case. The varactors are fed with the same DC bias voltage. Figure 18(b) shows the measured frequency response when the DC bias voltage is varied from 8.1V to 25V. This filter exhibits a finite frequency transmission zero at the high end of the passband with a center frequency tuning range of 100 MHz, between 825 and 925 MHz. In this case, the even-mode frequency is always higher than the odd-mode frequency when the filter is tuned [46].
Figure 18, (a) Fabricated tunable dual-mode filter using a single DC bias containing a finite frequency transmission zero at the high end of the passband. (b) Measured frequency response [35]. (Image copyright European Microwave Association, EuMA. Used with permission.)
One can also adopt another modification scheme, as shown in the literature [35] for a tunable dual-mode filter tuned with a single dc bias and having a transmission zero at a finite frequency at the low end of the passband.
Tunable four-pole dual-mode filter
Two or more dual-mode, open-loop resonators can be cascaded to construct a tunable filter with a higher order. For example, Figure 19 shows a fabricated quadrupole tunable filter of this type [36]. Each dual-mode open-loop resonator is loaded with three Infineon BB857 varactors. The entire filter is tuned with a single DC bias circuit. The measured results when the DC bias was varied over the range of 10.6 V to 34.0 V are plotted in Fig. 20 and compared to the simulation results with different varactor capacitance values loaded.
Figure 19, Fabricated quadrupole tunable dual-mode filter [36]. (Image copyright European Microwave Association, EuMA. Used with permission.)
Figure 20 shows the quasi-elliptic function response of an experimental four-pole tunable bandpass filter with a finite frequency transmission zero on each side of the passband. The transmission zero near the low end of the passband is associated with the even mode inherent to the first dual-mode resonator, while the transmission zero near the high end of the passband is associated with the even mode of the second dual-mode resonator. Therefore, as the even-mode frequency is tuned, the associated transmission zeros are shifted accordingly. For a given DC bias voltage range, one can tune the filter over a tuning range of 0.86-0.96 GHz.
Figure 20, Comparison of the measured performance of the tunable filter with the simulation results. (a) S21 and (b) S11 [36].
Conclusion
In this paper, several electronically reconfigurable or tunable microstrip line filters have been presented. By employing different electronic control techniques, including RF MEMS and ferroelectrics, comb filter structures have been widely used to develop tunable or reconfigurable filters, although the bandwidths of such filters are usually small. The UWB filter shown in this paper uses a p-i-n diode, but other switchable elements have been implemented, such as metal semiconductor field effect tube (MESFET) switches. mESFET switches have lower DC power consumption, but all have large nonlinear distortion. One can likewise consider the use of RF MEMS [13]-[14] or PETs [18].
BST varactors have been used in electrically tuned microstrip line bandstop filters with slotted wire grounding structures. These structures allow easy implementation of tuning elements and DC bias circuits with better isolation from the main RF signal path on the other side of the substrate. Ferroelectric thin film tuned devices such as BST varactors are attractive for higher frequency applications, although their losses need to be minimized.
We have demonstrated a dual-mode microstrip line open-loop resonator filter that can be tuned, or electronically reconfigured, by controlling the resonant frequencies of the odd and even modes in a simple way, since the two modes of operation are not coupled to each other.
In addition to semiconductor varactors, other types of varactors and techniques have been implemented, such as ferroelectric thin films and RF MEMS varactors. A similar tuning technique can be applied to higher order filters, which has been demonstrated in a tuned quadrupole dual-mode microstrip line open-loop resonator filter with a quasi-elliptic function response. The frequency tuning range can be increased by selecting the appropriate varactor diodes and by designing the input and output feed structures appropriately. It is also possible to tune the filter bandwidth by using additional tuning elements to control the input and output coupling.
In general, tuning or controlling the bandwidth is more challenging than tuning the frequency, and the design of ESC filters with larger bandwidths is more difficult in terms of tuning range and bandwidth control than narrow bandwidths. Some techniques for bandwidth control in tunable filters have been reported in the literature [48]-[51].
The nonlinear behavior of electronically reconfigurable filters is very dependent on the tuning elements used. The use of RF MEMS and PET usually produces a better linear characteristic. Innovations in reconfigurable filter design can also improve performance and can increase functionality. The relatively low Q of the tuning element can limit the implementation of higher order and narrow band tunable filters. This is because, for a given Q value of the tuned element and other losses associated with the circuit, the insertion loss of the filter increases as its order increases and the bandwidth decreases. Therefore, the insertion loss of a high-order tunable narrowband filter is too large for practical applications. Moreover, the tuning range is more limited in higher-order filters than in lower-order filters [4].
The development of reconfigurable filters involves some tradeoffs such as filter size and bias circuit complexity, which add to the challenge. It can be envisioned that there will be more research and development activities in developing electronically reconfigurable microstrip line filters. There are numerous papers on this topic, some of which are listed in the references, and the interested reader is referred to them for more information.
Acknowledgments
Some of the work presented in this paper was funded by the U.K. Engineering and Physical Science Research Council through two research projects (EP/C520289/1) and (EP/E02923/1). The authors would like to thank the organizations and individuals who participated in these projects.
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